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19-0723; Rev 2; 12/10
KIT ATION EVALU E AILABL AV
4.5V to 40V Input Automotive Flyback/Boost/SEPIC Power-Supply Controllers
General Description
The MAX15004A/B/MAX15005A/B high-performance, current-mode PWM controllers operate at an automotive input voltage range from 4.5V to 40V (load dump). The input voltage can go down as low as 2.5V after startup if VCC is supplied by an external bias voltage. The controllers integrate all the building blocks necessary for implementing fixed-frequency isolated/nonisolated power supplies. The general-purpose boost, flyback, forward, and SEPIC converters can be designed with ease around the MAX15004/MAX15005. The current-mode control architecture offers excellent line-transient response and cycle-by-cycle current limit while simplifying the frequency compensation. Programmable slope compensation simplifies the design further. A fast 60ns current-limit response time, low 300mV current-limit threshold makes the controllers suitable for high-efficiency, high-frequency DC-DC converters. The devices include an internal error amplifier and 1% accurate reference to facilitate the primary-side regulated, single-ended flyback converter or nonisolated converters. An external resistor and capacitor network programs the switching frequency from 15kHz to 500kHz (1MHz for the MAX15005A/B). The MAX15004A/B/MAX15005A/B provide a SYNC input for synchronization to an external clock. The maximum FET-driver duty cycle for the MAX15004A/B is 50%. The maximum duty cycle can be set on the MAX15005A/B by selecting the right combination of RT and CT. The input undervoltage lockout (ON/OFF) programs the input-supply startup voltage and can be used to shutdown the converter to reduce the total shutdown current down to 10A. Protection features include cycle-by-cycle and hiccup current limit, output overvoltage protection, and thermal shutdown. The MAX15004A/B/MAX15005A/B are available in space-saving 16-pin TSSOP and thermally enhanced 16-pin TSSOP-EP packages. All devices operate over the -40C to +125C automotive temperature range.
Features
o Wide 4.5V to 40V Operating Input Voltage Range o Operates Down to 2.5V (with Bootstrapped VCC Bias) o Current-Mode Control o Low 300mV, 5% Accurate Current-Limit Threshold Voltage o Internal Error Amplifier with 1% Accurate Reference o RC Programmable 4% Accurate Switching Frequency o Switching Frequency Adjustable from 15kHz to 500kHz (1MHz for the MAX15005A/B) o External Frequency Synchronization o 50% (MAX15004) or Adjustable (MAX15005) Maximum Duty Cycle o Programmable Slope Compensation o 10A Shutdown Current o Cycle-by-Cycle and Hiccup Current-Limit Protection o Overvoltage and Thermal Shutdown Protection o -40C to +125C Automotive Temperature Range o 16-Pin TSSOP or 16-Pin Thermally Enhanced TSSOP-EP Packages o AEC-Q100 Qualified
MAX15004/MAX15005
Ordering Information
PART MAX15004AAUE+ MAX15004BAUE+ MAX15005AAUE+ MAX15005BAUE+ PIN-PACKAGE 16 TSSOP-EP* 16 TSSOP 16 TSSOP-EP* 16 TSSOP MAX DUTY CYCLE 50% 50% Programmable Programmable
Applications
Automotive Vacuum Fluorescent Display (VFD) Power Supply Isolated Flyback, Forward, Nonisolated SEPIC, Boost Converters
MAX15005BAUE/V+ 16 TSSOP Programmable Note: All devices are specified over the -40C to +125C temperature range.
+Denotes lead (Pb)-free/RoHS-compliant package. *EP = Exposed pad.
/V = Automotive-qualified part. Pin Configurations appear at end of data sheet.
________________________________________________________________ Maxim Integrated Products
1
For pricing, delivery, and ordering information, please contact Maxim Direct at 1-888-629-4642, or visit Maxim's website at www.maxim-ic.com.
4.5V to 40V Input Automotive Flyback/Boost/SEPIC Power-Supply Controllers MAX15004/MAX15005
ABSOLUTE MAXIMUM RATINGS
IN to SGND.............................................................-0.3V to +45V IN to PGND.............................................................-0.3V to +45V ON/OFF to SGND ........................................-0.3V to (VIN + 0.3V) OVI, SLOPE, RTCT, SYNC, SS, FB, COMP, CS to SGND .........................................-0.3V to (VREG5 + 0.3V) VCC to PGND..........................................................-0.3V to +12V REG5 to SGND .........................................................-0.3V to +6V OUT to PGND .............................................-0.3V to (VCC + 0.3V) SGND to PGND .....................................................-0.3V to +0.3V VCC Sink Current (clamped mode) .....................................35mA OUT Current (< 10s transient) ..........................................1.5A Continuous Power Dissipation* (TA = +70C) 16-Pin TSSOP-EP (derate 21.3mW/C above +70C)..............................................................1702mW 16-Pin TSSOP (derate 9.4mW/C above +70C) ..........754mW Operating Junction Temperature Range ...........-40C to +125C Junction Temperature ......................................................+150C Storage Temperature Range .............................-60C to +150C Lead Temperature (soldering, 10s) .................................+300C Soldering Temperature (reflow) .......................................+260C *As per JEDEC51 Standard, Multilayer Board.
Stresses beyond those listed under "Absolute Maximum Ratings" may cause permanent damage to the device. These are stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to absolute maximum rating conditions for extended periods may affect device reliability.
ELECTRICAL CHARACTERISTICS
(VIN = 14V, CIN = 0.1F, CVCC = 0.1F // 1F, CREG5 = 1F, VON/OFF = 5V, CSS = 0.01F, CSLOPE = 100pF, RT = 13.7k, CT = 560pF, VSYNC = VOVI = VFB = VCS = 0V, COMP = unconnected, OUT = unconnected. TA = TJ = -40C to +125C, unless otherwise noted. Typical values are at TA = +25C. All voltages are referenced to PGND, unless otherwise noted.) (Note 1) (Figure 5)
PARAMETER POWER SUPPLY Input Supply Range Operating Supply Current ON/OFF CONTROL Input-Voltage Threshold Input-Voltage Hysteresis Input Bias Current Shutdown Current INTERNAL 7.4V LDO (VCC) Output (VCC) Voltage Set Point Line Regulation UVLO Threshold Voltage UVLO Hysteresis Dropout Voltage Output Current Limit Internal Clamp Voltage INTERNAL 5V LDO (REG5) Output (REG5) Voltage Set Point Line Regulation Dropout Voltage Output Current Limit IREG5-ILIM VREG5 VCC = 7.5V, IREG5 = 0 to 15mA (sourcing) VCC = 5.5V to 10V VCC = 4.5V, IREG5 = 15mA (sourcing) IREG5 sourcing 4.75 4.95 2 0.25 32 0.5 5.05 V mV/V V mA IVCC-ILIM VUVLO-VCC VHYST-UVLO VIN = 4.5V, IVCC = 20mA (sourcing) IVCC sourcing 10.0 VVCC-CLAMP IVCC = 30mA (sinking) VVCC IVCC = 0 to 20mA (sourcing) VIN = 8V to 40V VCC rising 3.15 7.15 7.4 1 3.5 500 0.25 45 10.4 10.8 0.5 3.75 7.60 V mV/V V mV V mA V VON VHYST-ON IB-ON/OFF ISHDN VON/OFF = 40V VON/OFF = 0V 10 VON/OFF rising 1.05 1.23 75 0.5 20 1.40 V mV A A VIN IQ VIN = 40V, fOSC = 150kHz 4.5 2 40.0 3.1 V mA SYMBOL CONDITIONS MIN TYP MAX UNITS
2
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4.5V to 40V Input Automotive Flyback/Boost/SEPIC Power-Supply Controllers
ELECTRICAL CHARACTERISTICS (continued)
(VIN = 14V, CIN = 0.1F, CVCC = 0.1F // 1F, CREG5 = 1F, VON/OFF = 5V, CSS = 0.01F, CSLOPE = 100pF, RT = 13.7k, CT = 560pF, VSYNC = VOVI = VFB = VCS = 0V, COMP = unconnected, OUT = unconnected. TA = TJ = -40C to +125C, unless otherwise noted. Typical values are at TA = +25C. All voltages are referenced to PGND, unless otherwise noted.) (Note 1) (Figure 5)
PARAMETER OSCILLATOR (RTCT) Oscillator Frequency Range RTCT Peak Trip Level RTCT Valley Trip Level RTCT Discharge Current fOSC VTH,RTCT VTL,RTCT IDIS,RTCT VRTCT = 2V RT = 13.7k, CT = 4.7nF, fOSC (typ) = 18kHz Oscillator Frequency Accuracy (Note 2) RT = 13.7k, CT = 560pF, fOSC (typ) = 150kHz RT = 21k, CT = 100pF, fOSC (typ) = 500kHz RT = 7k, CT = 100pF, fOSC (typ) = 1MHz MAX15004A/B Maximum PWM Duty Cycle (Note 3) Minimum On-Time SYNC Lock-In Frequency Range (Note 4) SYNC High-Level Voltage SYNC Low-Level Voltage SYNC Input Current SYNC Minimum Input Pulse Width ERROR AMPLIFIER/SOFT-START Soft-Start Charging Current SS Reference Voltage SS Threshold for HICCUP Enable FB Regulation Voltage VREF-FB ISS VSS VSS rising COMP = FB, ICOMP = -500A to +500A COMP = 0.25V to 4.5V, ICOMP = -500A to +500A, VSS = 0 to 1.5V VFB = 0 to 1.5V ICOMP-SINK VFB = 1.5V, VCOMP = 0.25V 1.215 VSS = 0V 8 1.215 15 1.228 1.1 1.228 1.240 21 1.240 A V V V VIH-SYNC VIL-SYNC ISYNC VSYNC = 0 to 5V -0.5 50 DMAX MAX15005A/B, RT = 13.7k, CT = 560pF, fOSC (typ) = 150kHz VIN = 14V RT = 13.7k, CT = 560pF, fOSC (typ) = 150kHz 102 2 0.8 +0.5 78.5 80 110 1.30 -4 -4 -5 -7 fOSC = 2 x fOUT for MAX15004A/B, fOSC = fOUT for MAX15005A/B 15 0.55 x VREG5 0.1 x VREG5 1.33 1.36 +4 +4 % +5 +7 50 81.5 170 200 % 1000 kHz V V mA SYMBOL CONDITIONS MIN TYP MAX UNITS
MAX15004/MAX15005
tON-MIN
ns %fOSC V V A ns
FB Input Offset Voltage FB Input Current COMP Sink Current
VOS-FB
-5 -300 3 5.5
+5 +300
mV nA mA
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4.5V to 40V Input Automotive Flyback/Boost/SEPIC Power-Supply Controllers MAX15004/MAX15005
ELECTRICAL CHARACTERISTICS (continued)
(VIN = 14V, CIN = 0.1F, CVCC = 0.1F // 1F, CREG5 = 1F, VON/OFF = 5V, CSS = 0.01F, CSLOPE = 100pF, RT = 13.7k, CT = 560pF, VSYNC = VOVI = VFB = VCS = 0V, COMP = unconnected, OUT = unconnected. TA = TJ = -40C to +125C, unless otherwise noted. Typical values are at TA = +25C. All voltages are referenced to PGND, unless otherwise noted.) (Note 1) (Figure 5)
PARAMETER COMP Source Current COMP High Voltage COMP Low Voltage Open-Loop Gain Unity-Gain Bandwidth Phase Margin COMP Positive Slew Rate COMP Negative Slew Rate PWM COMPARATOR Current-Sense Gain PWM Propagation Delay to OUT PWM Comparator Current-Sense Leading-Edge Blanking Time CURRENT-LIMIT COMPARATOR Current-Limit Threshold Voltage Current-Limit Input Bias Current ILIMIT Propagation Delay to OUT ILIM Comparator Current-Sense Leading-Edge Blanking Time Number of Consecutive ILIMIT Events to HICCUP HICCUP Timeout SLOPE COMPENSATION (Note 6) Slope Capacitor Charging Current Slope Compensation Slope Compensation Tolerance (Note 2) Slope Compensation Range ISLOPE VSLOPE = 100mV CSLOPE = 100pF CSLOPE = 100pF CSLOPE = 22pF CSLOPE = 1000pF -4 110 2.5 9.8 10.5 25 +4 11.2 A mV/s % mV/s VILIM IB-CS tPD-ILIM OUT= high, 0 VCS 0.3V From CS rising above VILIM (50mV overdrive) to OUT falling (excluding leading-edge blanking time) 290 -2 60 305 317 +2 mV A ns ACS-PWM tPD-PWM VCOMP/VCS (Note 5) CS = 0.15V, from VCOMP falling edge: 3V to 0.5V to OUT falling (excluding leading-edge blanking time) 2.85 3 60 3.15 V/V ns SYMBOL ICOMPSOURCE VOH-COMP VOL-COMP AEAMP UGFEAMP PMEAMP SR+ SRCONDITIONS VFB = 1V, VCOMP = 4.5V VFB = 1V, ICOMP = 1mA (sourcing) VFB = 1.5V, ICOMP = 1mA (sinking) MIN 1.3 VREG5 - 0.5 TYP 2.8 VREG5 - 0.2 0.1 100 1.6 75 0.5 -0.5 0.25 MAX UNITS mA V V dB MHz degrees V/s V/s
tCS-BLANK
50
ns
tCS-BLANK
50 7 512
ns
Clock periods
4
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4.5V to 40V Input Automotive Flyback/Boost/SEPIC Power-Supply Controllers
ELECTRICAL CHARACTERISTICS (continued)
(VIN = 14V, CIN = 0.1F, CVCC = 0.1F // 1F, CREG5 = 1F, VON/OFF = 5V, CSS = 0.01F, CSLOPE = 100pF, RT = 13.7k, CT = 560pF, VSYNC = VOVI = VFB = VCS = 0V, COMP = unconnected, OUT = unconnected. TA = TJ = -40C to +125C, unless otherwise noted. Typical values are at TA = +25C. All voltages are referenced to PGND, unless otherwise noted.) (Note 1) (Figure 5)
PARAMETER OUTPUT DRIVER ROUT-N Driver Output Impedance ROUT-P Driver Peak Output Current OVERVOLTAGE COMPARATOR Overvoltage Comparator Input Threshold Overvoltage Comparator Hysteresis Overvoltage Comparator Delay OVI Input Current THERMAL SHUTDOWN Shutdown Temperature Thermal Hysteresis TSHDN THYST Temperature rising 160 15
o o
MAX15004/MAX15005
SYMBOL
CONDITIONS VCC = 8V (applied externally), IOUT = 100mA (sinking) VCC = 8V (applied externally), IOUT = 100mA (sourcing) COUT = 10nF, sinking COUT = 10nF, sourcing
MIN
TYP
MAX
UNITS
1.7 3 1000 750
3.5 5 mA
IOUT-PEAK
VOV-TH VOV-HYST TDOVI IOVI
VOVI rising
1.20
1.228 125
1.26
V mV s
From OVI rising above 1.228V to OUT falling, with 50mV overdrive VOVI = 0 to 5V -0.5
1.6 +0.5
A C C
Note 1: 100% production tested at +125C. Limits over the temperature range are guaranteed by design. Note 2: Not production tested, guaranteed by design. Note 3: For the MAX15005A/B, DMAX depends upon the value of RT. See Figure 3 and the Oscillator Frequency/External Synchronization section. Note 4: The external SYNC pulse triggers the discharge of the oscillator ramp. See Figure 2. During external SYNC, DMAX = 50% for the MAX15004A/B; for the MAX15005A/B, there is a shift in DMAX with fSYNC/fOSC ratio (see the Oscillator Frequency/ External Synchronization section). Note 5: The parameter is measured at the trip point of latch, with 0 VCS 0.3V, and FB = COMP. Note 6: Slope compensation = (2.5 x 10-9)/CSLOPE mV/s. See the Applications Information section.
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4.5V to 40V Input Automotive Flyback/Boost/SEPIC Power-Supply Controllers MAX15004/MAX15005
Typical Operating Characteristics
(VIN = 14V, CIN = 0.1F, CVCC = 0.1F // 1F, CREG5 = 1F, VON/OFF = 5V, CSS = 0.01F, CSLOPE = 100pF, RT = 13.7k, CT = 560pF. TA = +25C, unless otherwise noted.)
VIN UVLO HYSTERESIS vs. TEMPERATURE
120 110 100 90 80 70 60 50 40 30 20 10 0 -40 -15 10 35 60 85 TEMPERATURE (C) 110 135
MAX15004 toc01
VIN SUPPLY CURRENT (ISUPPLY) vs. OSCILLATOR FREQUENCY (fOSC)
31 28 VIN SUPPLY CURRENT (mA) 25 22 19 16 13 10 7 4 1 10 60 110 160 210 260 310 360 410 460 510 FREQUENCY (kHz) COUT = 0nF COUT = 10nF VIN SHUTDOWN SUPPLY CURRENT (A) MAX15005 VIN = 14V CT = 220pF 20 19 18 17 16 15 14 13 12 11 10 9 8 7 6 5 5
MAX15004 toc02
SHUTDOWN SUPPLY CURRENT vs. SUPPLY VOLTAGE
MAX15004 toc03
VIN UVLO HYSTERESIS (mV)
TA = +135C
TA = +25C
TA = -40C 10 15 20 25 30 35 SUPPLY VOLTAGE (V) 40 45
VCC OUTPUT VOLTAGE vs. VIN SUPPLY VOLTAGE
MAX15004 toc04
VCC CLAMP VOLTAGE vs. VCC CURRENT SINK (IVCC)
10.50 10.25 10.00 9.75 9.50 9.25 9.00 8.75 8.50 8.25 8.00 7.75 7.50 7.25 7.00 TA = +25C TA = -40C 5.000 4.975 4.950 4.925 4.900 4.875 4.850 4.825 4.800 4.775 4.750 4.725 4.700
MAX15004 toc05
REG5 OUTPUT VOLTAGE vs. VCC VOLTAGE
IREG5 = 1mA (SOURCING)
MAX15004 toc06
7.5 IVCC = 0mA IVCC = 1mA
VCC OUTPUT VOLTAGE (V)
6.5
VCC CLAMP VOLTAGE (V)
IVCC = 20mA
REG5 OUTPUT VOLTAGE (V)
7.0
TA = +135C TA = +125C
IREG5 = 15mA (SOURCING)
6.0
5.5
5.0 5 10 15 20 25 30 35 VIN SUPPLY VOLTAGE (V) 40 45
0 2 4 6 8 10 12 14 16 18 20 22 24 26 28 30 VCC CURRENT SINK (mA)
5.5 6.0 6.5 7.0 7.5 8.0 8.5 9.0 9.5 10.0 10.5 VCC VOLTAGE (V)
REG5 DROPOUT VOLTAGE vs. IREG5
0.30 0.28 0.25 0.23 0.20 0.18 0.15 0.13 0.10 0.08 0.05 0.03 0 0 2 VCC = 4.5 VIN = VON/OFF
MAX15004 toc07
OSCILLATOR FREQUENCY (fOSC) vs. VIN SUPPLY VOLTAGE
MAX15004 toc08
OSCILLATOR FREQUENCY (fOSC) vs. RT/CT
CT = 100pF OSCILLATOR FREQUENCY (kHz) CT = 220pF CT = 560pF CT = 1000pF 100 CT = 1500pF CT = 2200pF CT = 3300pF 10
MAX15004 toc09
150 149 OSCILLATOR FREQUENCY (kHz) 148 147 146 145 144 143 142 141 140 TA = +125C TA = +135C TA = +25C TA = -40C RT = 13.7k CT = 560pF MAX15005
1000
TA = +125C
REG5 LDO DROPOUT VOLTAGE (V)
TA = +135C
TA = +25C TA = -40C 4 6 8 10 IREG5 (mA) 12 14
5.5 10.5 15.5 20.5 25.5 30.5 35.5 40.5 45.5 VIN SUPPLY VOLTAGE (V)
1
10 RT (k)
100
1000
6
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4.5V to 40V Input Automotive Flyback/Boost/SEPIC Power-Supply Controllers
Typical Operating Characteristics (continued)
(VIN = 14V, CIN = 0.1F, CVCC = 0.1F // 1F, CREG5 = 1F, VON/OFF = 5V, CSS = 0.01F, CSLOPE = 100pF, RT = 13.7k, CT = 560pF. TA = +25C, unless otherwise noted.)
MAX15005 MAXIMUM DUTY CYCLE vs. OUTPUT FREQUENCY (fOUT)
MAX15004 toc10
MAX15004/MAX15005
MAX15004 MAXIMUM DUTY CYCLE vs. TEMPERATURE
MAX15004 toc11
MAX15005 MAXIMUM DUTY CYCLE vs. TEMPERATURE
83 MAXIMUM DUTY CYCLE (%) 81 79 77 75 73 71 69 67 65 CT = 560pF RT = 13.7k fOSC = fOUT = 150kHz
MAX15004 toc12
100 95 MAXIMUM DUTY CYCLE (%) 90 85 80 75 70 65 60 55 50 10 100 OUTPUT FREQUENCY (kHz) CT = 1000pF CT = 560pF CT = 220pF CT = 3300pF CT = 2200pF CT = 1500pF CT = 100pF
55 54 MAXIMUM DUTY CYCLE (%) 53 52 51 50 49 48 47 46 45 fOUT = 75kHz
85
1000
-40
-15
10 35 60 85 TEMPERATURE (C)
110
135
-40
-15
10 35 60 85 TEMPERATURE (C)
110
135
MAXIMUM DUTY CYCLE vs. fSYNC/fOSC RATIO
80 MAXIMUM DUTY CYCLE (%) 75 70 GAIN (dB) 65 60 55 50 1.0 1.1 1.2 1.3 1.4 1.5 1.6 1.7 1.8 1.9 2.0 fSYNC/fOSC RATIO CRTCT = 220pF RRTCT = 10k fOSC = fOUT = 418kHz
MAX15004 toc13
ERROR AMPLIFIER OPEN-LOOP GAIN AND PHASE vs. FREQUENCY
110 100 90 80 70 60 50 40 30 20 10 0 -10 0.1 1 10 100 1k 10k 100k 1M 10M FREQUENCY (Hz) GAIN
MAX15004 toc14
CS-TO-OUT DELAY vs. TEMPERATURE
100 340 300 CS-TO-OUT DELAY (ns) PHASE (DEGREES) 260 220 180 90 80 70 60 50 40 30 20 100 60 10 0 -40 -15 10 35 60 85 TEMPERATURE (C) 110 135 VCS OVERDRIVE = 190mV VCS OVERDRIVE = 50mV
MAX15004 toc15
MAX15005 CT = 560pF RT = 10k fOSC = fOUT = 180kHz
PHASE
140
OVI TO OUT DELAY THROUGH OVERVOLTAGE COMPARATOR
MAX15004 toc16
DRIVER OUTPUT PEAK SOURCE AND SINK CURRENT
MAX15004 toc17
POWER-UP SEQUENCE THROUGH VIN
MAX15004 toc18
COUT = 10nF VOUT VOVI VOUT 5V/div VON/OFF = 5V VIN 10V/div VCC 5V/div REG5 5V/div
VOUT 2V/div
VOVI 500mV/div
IOUT 1A/div
VOUT 5V/div
1s/div
400ns/div
2ms/div
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4.5V to 40V Input Automotive Flyback/Boost/SEPIC Power-Supply Controllers MAX15004/MAX15005
Typical Operating Characteristics (continued)
(VIN = 14V, CIN = 0.1F, CVCC = 0.1F // 1F, CREG5 = 1F, VON/OFF = 5V, CSS = 0.01F, CSLOPE = 100pF, RT = 13.7k, CT = 560pF. TA = +25C, unless otherwise noted.)
POWER-UP SEQUENCE THROUGH ON/OFF
MAX15004 toc20
POWER-DOWN SEQUENCE THROUGH VIN
MAX15004 toc19
POWER-DOWN SEQUENCE THROUGH ON/OFF
MAX15004 toc21
VON/OFF = 5V VIN 10V/div VCC 5V/div REG5 5V/div VOUT 5V/div 4ms/div 1ms/div
ON/OFF 5V/div VCC 5V/div REG5 5V/div
ON/OFF 5V/div VCC 5V/div REG5 5V/div
VOUT 5V/div
VOUT 5V/div 400ms/div
LINE TRANSIENT FOR VIN STEP FROM 14V TO 5.5V
MAX15004 toc22
LINE TRANSIENT FOR VIN STEP FROM 14V TO 40V
MAX15004 toc23
VIN 10V/div VCC 5V/div REG5 5V/div
VIN 20V/div VCC 5V/div REG5 5V/div
VOUT 5V/div
VOUT 5V/div
100s/div
100s/div
HICCUP MODE FOR FLYBACK CIRCUIT (FIGURE 7)
MAX15004 toc24
DRAIN WAVEFORM IN FLYBACK CONVERTER (FIGURE 7)
MAX15004 toc25
ILOAD = 10mA VCS 200mV/div 10V/div VANODE 1V/div
ISHORT 500mA/div
10ms/div
4s/div
8
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4.5V to 40V Input Automotive Flyback/Boost/SEPIC Power-Supply Controllers
Pin Description
PIN 1 2 NAME IN ON/OFF FUNCTION Input Power Supply. Bypass IN with a minimum 0.1F ceramic capacitor to PGND. ON/OFF Input. Connect ON/OFF to IN for always-on operation. To externally program the UVLO threshold of the IN supply, connect a resistive divider between IN, ON/OFF, and SGND. Pull ON/OFF to SGND to disable the controller. Overvoltage Comparator Input. Connect a resistive divider between the output of the power supply, OVI, and SGND to set the output overvoltage threshold. Programmable Slope Compensation Capacitor Input. Connect a capacitor (CSLOPE) to SGND to set the amount of slope compensation. Slope compensation = (2.5 x 10-9) / CSLOPE mV/s with CSLOPE in farads. No Connection. Not internally connected. Oscillator-Timing Network Input. Connect a resistor from RTCT to REG5 and a capacitor from RTCT to SGND to set the oscillator frequency (see the Oscillator Frequency/External Synchronization section). Signal Ground. Connect SGND to SGND plane. External-Clock Synchronization Input. Connect SYNC to SGND when not using an external clock. Soft-Start Capacitor Input. Connect a capacitor from SS to SGND to set the soft-start time interval. Internal Error-Amplifier Inverting Input. The noninverting input is internally connected to SS. Error-Amplifier Output. Connect the frequency compensation network between FB and COMP. Current-Sense Input. The current-sense signal is compared to a signal proportional to the error-amplifier output voltage. 5V Low-Dropout Regulator Output. Bypass REG5 with a 1F ceramic capacitor to SGND. Power Ground. Connect PGND to the power ground plane. Gate Driver Output. Connect OUT to the gate of the external n-channel MOSFET. 7.4V Low-Dropout Regulator Output--Driver Power Source. Bypass VCC with 0.1F and 1F or higher ceramic capacitors to PGND. Exposed Pad (MAX15004A/MAX15005A only). Connect EP to the SGND plane to improve thermal performance. Do not use the EP as an electrical connection.
MAX15004/MAX15005
3
OVI
4 5 6 7 8 9 10 11 12 13 14 15 16 --
SLOPE N.C. RTCT SGND SYNC SS FB COMP CS REG5 PGND OUT VCC EP
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9
4.5V to 40V Input Automotive Flyback/Boost/SEPIC Power-Supply Controllers MAX15004/MAX15005
Functional Diagram
IN 1
MAX15004A/B MAX15005A/B
1.228V PREREGULATOR OFF ON/OFF COMP OFF 7.4V LDO REG 10.5V 30mA CLAMP REFERENCE 3.5V UVLO UVB DRIVER 15 OUT 16 VCC
ON/OFF 2
14 PGND THERMAL SHUTDOWN SET UVB VCC
5V LDO REG
13 REG5
OVI 3 1.228V
OV-COMP
RESET
ILIMIT COMP
0.3V 50ns LEAD DELAY R 12 CS
PWMCOMP OVRLD SLOPE 4 SLOPE COMPENSATION 2R
RTCT 6
OSCILLATOR CLK
SS_OK EAMP 1.228V
11 COMP 10 FB
SGND 7 RESET SYNC 8 7 CONSECUTIVE EVENTS COUNTER
9 SS
REF-AMP OVRLD
10
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4.5V to 40V Input Automotive Flyback/Boost/SEPIC Power-Supply Controllers
Detailed Description
The MAX15004A/B/MAX15005A/B are high-performance, current-mode PWM controllers for wide inputvoltage range isolated/nonisolated power supplies. These controllers are for use as general-purpose boost, flyback, and SEPIC controllers. The input voltage range of 4.5V to 40V makes it ideal in automotive applications such as vacuum fluorescent display (VFD) power supplies. The internal low-dropout regulator (VCC regulator) enables the MAX15004A/B/MAX15005A/B to operate directly from an automotive battery input. The input operating range can be as low as 2.5V when an external source (e.g. bootstrap winding output) is applied at the VCC input. The 2.5V to 40V input voltage range allows device operation from cold crank to automotive load dump. The undervoltage lockout (ON/OFF) allows the devices to program the input-supply startup voltage and ensures predictable operation during brownout conditions. The devices contain two internal regulators, VCC and REG5. The VCC regulator output voltage is set at 7.4V and REG5 regulator output voltage at 5V 2%. The VCC output includes a 10.4V clamp that is capable of sinking up to 30mA current. The input undervoltage lockout (UVLO) circuit monitors the VCC voltage and turns off the converter when the VCC voltage drops below 3.5V (typ). See the Internal Regulators VCC and REG5 section for a method to obtain lower than 4.5V input operation with the MAX15004/MAX15005. An external resistor and capacitor network programs the switching frequency from 15kHz to 500kHz. The MAX15004A/B/MAX15005A/B provide a SYNC input for synchronization to an external clock. The OUT (FET-driver output) duty cycle for the MAX15004A/B is 50%. The maximum duty cycle can be set on MAX15005A/B by selecting the right combination of RT and CT. The RTCT discharge current is trimmed to 2%, allowing accurate setting of the duty cycle for the MAX15005. An internal slope-compensation circuit stabilizes the current loop when operating at higher duty cycles and can be programmed externally. The MAX15004/MAX15005 include an internal error amplifier with 1% accurate reference to regulate the output in nonisolated topologies using a resistive divider. The internal reference connected to the noninverting input of the error amplifier can be increased in a controlled manner to obtain soft-start. A capacitor connected at SS to ground programs soft-start to reduce inrush current and prevent output overshoot. The MAX15004/MAX15005 include protection features like hiccup current limit, output overvoltage, and thermal shutdown. The hiccup current-limit circuit reduces the power delivered to the electronics powered by the MAX15004/MAX15005 converter during severe fault conditions. The overvoltage circuit senses the output using the path different from the feedback path to provide meaningful overvoltage protection. During continuous high input operation, the power dissipation into the MAX15004/MAX15005 could exceed its limit. Internal thermal shutdown protection safely turns off the converter when the junction heats up to 160C.
MAX15004/MAX15005
Current-Mode Control Loop
The advantages of current-mode control overvoltagemode control are twofold. First, there is the feed-forward characteristic brought on by the controller's ability to adjust for variations in the input voltage on a cycleby-cycle basis. Secondly, the stability requirements of the current-mode controller are reduced to that of a single-pole system unlike the double pole in voltage-mode control. The MAX15004/MAX15005 offer peak current-mode control operation to make the power supply easy to design with. The inherent feed-forward characteristic is useful especially in an automotive application where the input voltage changes fast during cold-crank and load dump conditions. While the current-mode architecture offers many advantages, there are some shortcomings. For higher duty-cycle and continuous conduction mode operation where the transformer does not discharge during the off duty cycle, subharmonic oscillations appear. The MAX15004/MAX15005 offer programmable slope compensation using a single capacitor. Another issue is noise due to turn-on of the primary switch that may cause the premature end of the on cycle. The current-limit and PWM comparator inputs have leadingedge blanking. All the shortcomings of the current-mode control are addressed in the MAX15004/ MAX15005, making it ideal to design for automotive power conversion applications.
Internal Regulators VCC and REG5 The internal LDO converts the automotive battery voltage input to a 7.4V output voltage (VCC). The VCC output is set at 7.4V and operates in a dropout mode at input voltages below 7.5V. The internal LDO is capable of delivering 20mA current, enough to provide power to internal control circuitry and the gate drive. The regulated VCC keeps the driver output voltage well below the absolute maximum gate voltage rating of the MOSFET especially during the double battery and load dump conditions. An auxiliary winding output can be fed to the VCC output once the power supply is turned on. The bootstrap winding is not necessary for proper
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4.5V to 40V Input Automotive Flyback/Boost/SEPIC Power-Supply Controllers MAX15004/MAX15005
operation of the power supply; however, to reduce the power dissipation of the internal LDO, it can be disabled by applying an external voltage higher than 7.4V at VCC (LDO output). The LDO then stops drawing current from IN, thereby reducing the power dissipation in the IC. The VCC voltage is clamped to 10.4V with 30mA current sink in case there is a higher voltage at the bias winding. This feature is useful in applications with continuous higher input voltage. The second 5V LDO regulator from VCC to REG5 provides power to the internal control circuits. This LDO can also be used to source 15mA of external load current. Bypass VCC and REG5 with a parallel combination of 1F and 0.1F low-ESR ceramic capacitors. Additional capacitors (up to 22F) at VCC can be used although they are not necessary for proper operation of the MAX15004/MAX15005.
VIN
MAX15004A/B MAX15005A/B
ON/OFF 1.23V
R1
R2
Figure 1. Setting the MAX15004A/B/MAX15005A/B Undervoltage Lockout Threshold
Startup Operation/UVLO/ON/OFF
The MAX15004A/B/MAX15005A/B feature two undervoltage lockouts (UVLO). The internal UVLO monitors the VCC-regulator and turns on the converter once VCC rises above 3.5V. The internal UVLO circuit has about 0.5V hysteresis to avoid chattering during turn-on. Once the power is on and the bootstrapped voltage feeds VCC, IN voltage can drop below 4V. This feature provides operation at a cold-crank voltage as low as 2.5V. An external undervoltage lockout can be achieved by controlling the voltage at the ON/OFF input. The ON/OFF input threshold is set at 1.23V (rising) with 75mV hysteresis. Before any operation can commence, the ON/OFF voltage must exceed the 1.23V threshold. Calculate R1 in Figure 1 by using the following formula: V R1 = ON - 1 x R2 VUVLO where VUVLO is the ON/OFF's 1.23V rising threshold, and VON is the desired input startup voltage. Choose an R2 value in the 100k range. The UVLO circuits keep the PWM comparator, ILIM comparator, oscillator, and output driver shut down to reduce current consumption (see the Functional Diagram). The ON/OFF input can be used to disable the MAX15004/MAX15005 and reduce the standby current to less than 20A.
and also controls the soft-start period. At startup, after VIN is applied and the UVLO thresholds are reached, the device enters soft-start. During soft-start, 15A is sourced into the capacitor (CSS) connected from SS to GND causing the reference voltage to ramp up slowly. The HICCUP mode of operation is disabled during softstart. When VSS reaches 1.228V, the output as well as the HICCUP mode become fully active. Set the soft-start time (tSS) using following equation: t SS = 1.23(V) x CSS 15 x 10-6 (A )
where tSS is in seconds and CSS is in farads. The soft-start programmability is important to control the input inrush current issue and also to avoid the MAX15004/MAX15005 power supply from going into the unintentional hiccup during the startup. The required soft-start time depends on the topology used, currentlimit setting, output capacitance, and the load condition.
Oscillator Frequency/ External Synchronization
Use an external resistor and capacitor at RTCT to program the MAX15004A/B/MAX15005A/B internal oscillator frequency from 15kHz to 1MHz. The MAX15004A/B output switching frequency is one-half the programmed oscillator frequency with a 50% maximum duty-cycle limit. The MAX15005A/B output switching frequency is the same as the oscillator frequency. The RC network connected to RTCT controls both the oscillator frequency and the maximum duty cycle. The CT capacitor charges and discharges from (0.1 x VREG5) to (0.55 x VREG5). It charges through RT and discharges through an internal trimmed controlled current sink. The maximum duty cycle is inversely proportional to the discharge time
Soft-Start
The MAX15004/MAX15005 are provided with an externally adjustable soft-start function, saving a number of external components. The SS is a 1.228V reference bypass connection for the MAX15004A/B/MAX15005A/B
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4.5V to 40V Input Automotive Flyback/Boost/SEPIC Power-Supply Controllers
(tDISCHARGE). See Figures 3a and 3b for a coarse selection of capacitor values for a given switching frequency and maximum duty cycle and then use the following equations to calculate the resistor value to fine-tune the switching frequency and verify the worst-case maximum duty cycle.
t CHARGE = DMAX fOSC
The MAX15004A/B is a 50% maximum duty-cycle part, while the MAX15005A/B is 100% maximum duty-cycle part. fOUT = for the MAX15004A/B and fOUT = fOSC for the MAX15005A/B. The MAX15004A/B/MAX15005A/B can be synchronized using an external clock at the SYNC input. For proper frequency synchronization, SYNC's input frequency must be at least 102% of the programmed internal oscillator frequency. Connect SYNC to SGND when not using an external clock. A rising clock edge on SYNC is interpreted as a synchronization input. If the SYNC signal is lost, the internal oscillator takes control of the switching rate, returning the switching frequency to that set by RC network connected to RTCT. This maintains output regulation even with intermittent SYNC signals.
WITHOUT SYNC INPUT
MAX15004/MAX15005
1 fOSC 2
tDISCHARGE =
t RT = CHARGE 0.7 x CT 2.25(V) x RT x CT (1.33 x 10 -3 (A) x RT) - 3.375(V)
1 ...................Use This Equation If fOSC 500kHz t CHARGE + tDISCHARGE fOSC = 1 .......Use This Equation If fOSC > 500kHz t CHARGE + tDISCHARGE + 160ns
where fOSC is the oscillator frequency, RT is a resistor connected from RTCT to REG5, and CT is a capacitor connected from RTCT to SGND. Verify that the oscillator frequency value meets the target. Above calculations could be repeated to fine-tune the switching frequency.
MAX15004A/B (DMAX = 50%)
WITH SYNC INPUT
RTCT CLKINT SYNC OUT D = 50% D = 50%
MAX15005A/B (DMAX = 81%)
WITH SYNC INPUT
WITHOUT SYNC INPUT
RTCT CLKINT SYNC OUT D = 81.25% D = 80%
Figure 2. Timing Diagram for Internal Oscillator vs. External SYNC and DMAX Behavior
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4.5V to 40V Input Automotive Flyback/Boost/SEPIC Power-Supply Controllers MAX15004/MAX15005
MAX15005 MAXIMUM DUTY CYCLE vs. OUTPUT FREQUENCY (fOUT)
100 95 MAXIMUM DUTY CYCLE (%) 90 85 80 75 70 65 60 55 50 10 100 OUTPUT FREQUENCY (kHz) 1000 CT = 1000pF CT = 560pF CT = 220pF 10 1 10 RT (k) 100 1000 CT = 3300pF CT = 2200pF CT = 1500pF CT = 100pF OSCILLATOR FREQUENCY (kHz) 1000 CT = 100pF CT = 220pF CT = 560pF CT = 1000pF 100 CT = 1500pF CT = 2200pF CT = 3300pF
OSCILLATOR FREQUENCY (fOSC) vs. RT/CT
Figure 3a. MAX15005 Maximum Duty Cycle vs. Output Frequency.
Figure 3b. Oscillator Frequency vs. RT/CT
n-Channel MOSFET Driver
OUT drives the gate of an external n-channel MOSFET. The driver is powered by the internal regulator (VCC), internally set to approximately 7.4V. If an external voltage higher than 7.4V is applied at VCC (up to 10V), it appears as the peak gate drive voltage. The regulated VCC voltage keeps the OUT voltage below the maximum gate voltage rating of the external MOSFET. OUT can source 750mA and sink 1000mA peak current. The average current sourced by OUT depends on the switching frequency and total gate charge of the external MOSFET.
Slope Compensation
The MAX15004A/B/MAX15005A/B use an internal ramp generator for slope compensation. The internal ramp signal resets at the beginning of each cycle and slews at the rate programmed by the external capacitor connected to SLOPE. The amount of slope compensation needed depends on the downslope of the current waveform. Adjust the MAX15004A/B/MAX15005A/B slew rate up to 110mV/s using the following equation: Slope compensation (mV s) = 2.5 x 10 -9 (A) CSLOPE
Error Amplifier
The MAX15004A/B/MAX15005A/B include an internal error amplifier. The noninverting input of the error amplifier is connected to the internal 1.228V reference and feedback is provided at the inverting input. High 100dB open-loop gain and 1.6MHz unity-gain bandwidth allow good closed-loop bandwidth and transient response. Moreover, the source and sink current capability of 2mA provides fast error correction during the output load transient. For Figure 5, calculate the powersupply output voltage using the following equation: R VOUT = 1 + A VREF RB where VREF = 1.228V. The amplifier's noninverting input is internally connected to a soft-start circuit that gradually increases the reference voltage during startup. This forces the output voltage to come up in an orderly and well-defined manner under all load conditions.
where CSLOPE is the external capacitor at SLOPE in farads.
Current Limit
The current-sense resistor (RCS), connected between the source of the MOSFET and ground, sets the current limit. The CS input has a voltage trip level (V CS) of 305mV. The current-sense threshold has 5% accuracy. Set the current-limit threshold 20% higher than the peak switch current at the rated output power and minimum input voltage. Use the following equation to calculate the value of RS: V RS = CS IPRI where IPRI is the peak current that flows through the MOSFET at full load and minimum VIN.
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4.5V to 40V Input Automotive Flyback/Boost/SEPIC Power-Supply Controllers
When the voltage produced by this current (through the current-sense resistor) exceeds the current-limit comparator threshold, the MOSFET driver (OUT) quickly terminates the on-cycle. In most cases, a short-time constant RC filter is required to filter out the leadingedge spike on the sense waveform. The amplitude and width of the leading edge depends on the gate capacitance, drain capacitance (including interwinding capacitance), and switching speed (MOSFET turn-on time). Set the RC time constant just long enough to suppress the leading edge. For a given design, measure the leading spike at the highest input and rated output load to determine the value of the RC filter. The low 305mV current-limit threshold reduces the power dissipation in the current-sense resistor. The current-limit threshold can be further reduced by adding an offset to the CS input from REG5 voltage. Do not reduce the current-limit threshold below 150mV as it may cause noise issues. See Figure 4. For a new value of the current-limit threshold (VCS-LOW), calculate the value of R1 using the following equation. R1 = 4.75 x RCS 0.290 - VCS - Low where:
VIN
Applications Information
Boost Converter
The MAX15004A/B/MAX15005A/B can be configured for step-up conversion. The boost converter output can be fed back to VCC (see Figure 5) so that the controller can function even during cold-crank input voltage ( 2.5V). Use a Schottky diode (DVIN) in the VIN path to avoid backfeeding the input source. A current-limiting resistor (RVCC) is also needed from the boost converter output to VCC depending upon the boost converter output voltage. The total current sink into VCC must be limited to 30mA. Use the equations in the following sections to calculate R VCC , inductor (L MIN ), input capacitor (C IN), and output capacitor (C OUT) when using the converter in boost operation.
MAX15004/MAX15005
Inductor Selection in Boost Configuration Using the following equation, calculate the minimum inductor value so that the converter remains in continuous mode operation at minimum output current (IOMIN).
LMIN = VIN2 x D x 2 x fOUT x VOUT x IOMIN
D= and
VOUT + VD - VIN VOUT + VD - VDS
REG5
IOMIN = (0.1x IO ) to (0.25 x IO )
MAX15004A/B MAX15005A/B
R1 RCS N
0.3V CURRENT-LIMIT COMPARATOR CCS
RS
The higher value of IOMIN reduces the required inductance; however, it increases the peak and RMS currents in the switching MOSFET and inductor. Use IOMIN from 10% to 25% of the full load current. The VD is the forward voltage drop of the external Schottky diode, D is the duty cycle, and VDS is the voltage drop across the external switch. Select the inductor with low DC resistance and with a saturation current (ISAT) rating higher than the peak switch current limit of the converter.
Figure 4. Reducing Current-Sense Threshold
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4.5V to 40V Input Automotive Flyback/Boost/SEPIC Power-Supply Controllers MAX15004/MAX15005
VIN DVIN CIN CREG5 0.1F 13 RT REG5 1 IN VCC 16 CVCC 4.7F 6 CT RTCT CVIN 1F RVCC L VOUT 18V COUT
D3 DVCC
MAX15004A/B MAX15005A/B
OUT
15
Q
CFF CS RF CF 11 10 COMP FB SLOPE 4 CSLOPE SS 9 CSS
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RCS RA CCS RS
PGND
RB
Figure 5. Application Schematic
Input Capacitor Selection in Boost Configuration The input current for the boost converter is continuous and the RMS ripple current at the input capacitor is low. Calculate the minimum input capacitor value and maximum ESR using the following equations:
IL x D 4 x fOUT x VQ V ESR = ESR IL CIN = where: IL = (VIN - VDS ) x D L x fOUT
capacitor discharge and VESR is the contribution due to ESR of the capacitor. Assume the input capacitor ripple contribution due to ESR (VESR) and capacitor discharge (VQ) is equal when using a combination of ceramic and aluminum capacitors. During the converter turn-on, a large current is drawn from the input source especially at high output to input differential. The MAX15004/MAX15005 are provided with a programmable soft-start, however, a large storage capacitor at the input may be necessary to avoid chattering due to finite hysteresis.
VDS is the total voltage drop across the external MOSFET plus the voltage drop across the inductor ESR. IL is peak-to-peak inductor ripple current as calculated above. VQ is the portion of input ripple due to the
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Output Capacitor Selection in Boost Configuration For the boost converter, the output capacitor supplies the load current when the main switch is on. The required output capacitance is high, especially at higher duty cycles. Also, the output capacitor ESR needs to be low enough to minimize the voltage drop due to the ESR while supporting the load current. Use the following equations to calculate the output capacitor, for a specified output ripple. All ripple values are peak-to-peak.
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4.5V to 40V Input Automotive Flyback/Boost/SEPIC Power-Supply Controllers
V ESR = ESR IO I x DMAX COUT = O VQ x fOUT IO is the load current, VQ is the portion of the ripple due to the capacitor discharge, and VESR is the contribution due to the ESR of the capacitor. DMAX is the maximum duty cycle at the minimum input voltage. Use a combination of low-ESR ceramic and high-value, low-cost aluminum capacitors for lower output ripple and noise. 2) Reverse-transfer capacitance or charge (CRSS). 3) On-resistance (RDS(ON)). 4) Maximum drain-to-source voltage (VDS(MAX)). 5) Maximum gate frequencies threshold voltage (VTH(MAX)). At high switching, dynamic characteristics (parameters 1 and 2 of the above list) that predict switching losses have more impact on efficiency than RDS(ON), which predicts DC losses. Qg includes all capacitances associated with charging the gate. The VDS(MAX) of the selected MOSFET must be greater than the maximum output voltage setting plus a diode drop. The 10V additional margin is recommended for spikes at the MOSFET drain due to the inductance in the rectifier diode and output capacitor path. In addition, Qg helps predict the current needed to drive the gate at the selected operating frequency when the internal LDO is driving the MOSFET.
MAX15004/MAX15005
Calculating Power Loss in Boost Converter The MAX15004A/MAX15005A devices are available in a thermally enhanced package and can dissipate up to 1.7W at +70C ambient temperature. The total power dissipation in the package must be limited so that the junction temperature does not exceed its absolute maximum rating of +150C at maximum ambient temperature; however, Maxim recommends operating the junction at about +125C for better reliability.
The average supply current (IDRIVE-GATE) required by the switch driver is: IDRIVE-GATE = Qg x fOUT where Qg is total gate charge at 7.4V, a number available from MOSFET datasheet. The supply current in the MAX15004A/B/MAX15005A/B is dependent on the switching frequency. See the Typical Operating Characteristics to find the supply current ISUPPLY of the MAX15004A/B/MAX15005A/B at a given operating frequency. The total power dissipation (PT) in the device due to supply current (ISUPPLY) and the current required to drive the switch (IDRIVEGATE) is calculated using following equation. PT = VINMAX x (ISUPPLY + IDRIVE - GATE )
Slope Compensation in Boost Configuration The MAX15004A/B/MAX15005A/B use an internal ramp generator for slope compensation to stabilize the current loop when operating at duty cycles above 50%. It is advisable to add some slope compensation even at lower than 50% duty cycle to improve the noise immunity. The slope compensations should be optimized because too much slope compensation can turn the converter into the voltage-mode control. The amount of slope compensation required depends on the downslope of the inductor current when the main switch is off. The inductor downslope depends on the input to output voltage differential of the boost converter, inductor value, and the switching frequency. Theoretically, the compensation slope should be equal to 50% of the inductor downslope; however, a little higher than 50% slope is advised. Use the following equation to calculate the required compensating slope (mc) for the boost converter:
mc = (VOUT - VIN ) x RS x 10 -3 (mV s) 2L
MOSFET Selection in Boost Converter The MAX15004A/B/MAX15005A/B drive a wide variety of n-channel power MOSFETs. Since VCC limits the OUT output peak gate-drive voltage to no more than 11V, a 12V (max) gate voltage-rated MOSFET can be used without an additional clamp. Best performance, especially at low-input voltages (5VIN), is achieved with low-threshold n-channel MOSFETs that specify on-resistance with a gate-source voltage (VGS) of 2.5V or less. When selecting the MOSFET, key parameters can include:
1) Total gate charge (Qg).
The internal ramp signal resets at the beginning of each cycle and slews at the rate programmed by the external capacitor connected to SLOPE. Adjust the MAX15004A/B/MAX15005A/B slew rate up to 110mV/s using the following equation: CSLOPE = 2.5 x 10 -9 mc(mV s)
where CSLOPE is the external capacitor at SLOPE in farads.
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4.5V to 40V Input Automotive Flyback/Boost/SEPIC Power-Supply Controllers MAX15004/MAX15005
Selecting VCC Resistor (RVCC) The VCC external supply series resistor should be sized to provide enough average current from VOUT to drive the external MOSFET (IDRIVE) and ISUPPLY. The VCC is clamped internally to 10.4V and capable of sinking 30mA current. The VCC resistor must be high enough to limit the VCC sink current below 30mA at the highest output voltage. Maintain the VCC voltage to 8V while feeding the power from VOUT to VCC. For a regulated output voltage of VOUT, calculate the RVCC using the following equation:
RVCC = (VOUT - 8) (ISUPPLY + IDRIVE ) able with a discontinuous mode flyback topology using the MAX15004/MAX15005 in automotive applications.
Transformer Design Step-by-step transformer specification design for a discontinuous flyback example is explained below.
Follow the steps below for the discontinuous mode transformer: Step 1) Calculate the secondary winding inductance for guaranteed core discharge within a minimum off-time. Step 2) Calculate primary winding inductance for sufficient energy to support the maximum load. Step 3) Calculate the secondary and bias winding turns ratios. Step 4) Calculate the RMS current in the primary and estimate the secondary RMS current. Step 5) Consider proper sequencing of windings and transformer construction for low leakage. Step 1) As discussed earlier, the core must be discharged during the off-cycle for discontinuous mode operation. The secondary inductance determines the time required to discharge the core. Use the following equations to calculate the secondary inductance: LS
See Figure 5 and the Power Dissipation section for the values of ISUPPLY and IDRIVE.
Flyback Converter
The choice of the conversion topology is the first stage in power-supply design. The topology selection criteria include input voltage range, output voltage, peak currents in the primary and secondary circuits, efficiency, form factor, and cost. For an output power of less than 50W and a 1:2 input voltage range with small form factor requirements, the flyback topology is the best choice. It uses a minimum of components, thereby reducing cost and form factor. The flyback converter can be designed to operate either in continuous or discontinuous mode of operation. In discontinuous mode of operation, the transformer core completes its energy transfer during the off-cycle, while in continuous mode of operation, the next cycle begins before the energy transfer is complete. The discontinuous mode of operation is chosen for the present example for the following reasons: * It maximizes the energy storage in the magnetic component, thereby reducing size. * Simplifies the dynamic stability compensation design (no right-half plane zero). * Higher unity-gain bandwidth. A major disadvantage of discontinuous mode operation is the higher peak-to-average current ratio in the primary and secondary circuits. Higher peak-to-average current means higher RMS current, and therefore, higher loss and lower efficiency. For low-power converters, the advantages of using discontinuous mode easily surpass the possible disadvantages. Moreover, the drive capability of the MAX15004/MAX15005 is good enough to drive a large switching MOSFET. With the presently available MOSFETs, power output of up to 50W is easily achiev18
(VOUT + VD ) x (DOFFMIN )2
2 x IOUT x fOUT(MAX) t OFF t ON + t OFF
DOFF =
where: DOFFMIN = minimum DOFF. VD = secondary diode forward voltage drop. IOUT = maximum output rated current. Step 2) The rising current in the primary builds the energy stored in the core during on-time, which is then released to deliver the output power during the off-time. Primary inductance is then calculated to store enough energy during the on-time to support the maximum output power.
2 V x DMAX 2 x LP = INMIN 2 x POUT x fOUT(MAX)
D=
t ON t ON + t OFF
DMAX = Maximum D.
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4.5V to 40V Input Automotive Flyback/Boost/SEPIC Power-Supply Controllers
Step 3) Calculate the secondary to primary turns ratio (NSP) and the bias winding to primary turns ratio (NBP) using the following equations: N LS NSP = S = NP LP and 11.7 N NBP = BIAS = NP VOUT + 0.35 The forward bias drops of the secondary diode and the bias rectifier diode are assumed to be 0.35V and 0.7V, respectively. Refer to the diode manufacturer's datasheet to verify these numbers. Step 4) The transformer manufacturer needs the RMS current maximum values in the primary, secondary, and bias windings to design the wire diameter for the different windings. Use only wires with a diameter smaller than 28AWG to keep skin effect losses under control. To achieve the required copper cross-section, multiple wires must be used in parallel. Multifilar windings are common in high-frequency converters. Maximum RMS currents in the primary and secondary occur at 50% duty cycle (minimum input voltage) and maximum output power. Use the following equations to calculate the primary and secondary RMS currents: IPRMS = ISRMS = POUT DMAX x 0.5 x DMAX x x VINMIN 3 IOUT 0.5 x DOFFMAX DOFFMAX 3 spike. The MOSFET's absolute maximum VDS rating must be higher than the worst-case (maximum input voltage and output load) drain voltage. N VDSMAX = VINMAX + P x (VOUT + VD ) + VSPIKE NS Lower maximum V DS requirement means a shorter channel, lower RDS-ON, lower gate charge, and smaller package. A lower NP/NS ratio allows a low VDSMAX specification and keeps the leakage inductance spike under control. A resistor/diode/capacitor snubber network can be also used to suppress the leakage inductance spike. The DC losses in the MOSFET can be calculated using the value for the primary RMS maximum current. Switching losses in the MOSFET depend on the operating frequency, total gate charge, and the transition loss during turn-off. There are no transition losses during turn-on since the primary current starts from zero in the discontinuous conduction mode. MOSFET derating may be necessary to avoid damage during system turn-on and any other fault conditions. Use the following equation to estimate the power dissipation due to the power MOSFET: PMOS = (1.4 x RDSON x I2PRMS ) + (Qg x VIN x fOUTMAX ) + V xI x t xf ( INMAX PK OFF OUTMAX ) 4 C x V 2 x fOUTMAX + DS DS 2 where: Qg = Total gate charge of the MOSFET (C) at 7.4V VIN = Input voltage (V) tOFF = Turn-off time (s) CDS = Drain-to-source capacitance (F)
MAX15004/MAX15005
The bias current for most MAX15004/MAX15005 applications is about 20mA and the selection of wire depends more on convenience than on current capacity. Step 5) The winding technique and the windings sequence is important to reduce the leakage inductance spike at switch turn-off. For example, interleave the secondary between two primary halves. Keep the bias winding close to the secondary, so that the bias voltage tracks the output voltage.
MOSFET Selection MOSFET selection criteria include the maximum drain voltage, peak/RMS current in the primary and the maximum-allowable power dissipation of the package without exceeding the junction temperature limits. The voltage seen by the MOSFET drain is the sum of the input voltage, the reflected secondary voltage through transformer turns ratio and the leakage inductance
Output Filter Design The output capacitance requirements for the flyback converter depend on the peak-to-peak ripple acceptable at the load. The output capacitor supports the load current during the switch on-time. During the off-cycle, the transformer secondary discharges the core replenishing the lost charge and simultaneously supplies the load current. The output ripple is the sum of the voltage drop due to charge loss during the switch on-time and the ESR of the output capacitor. The high switching frequency of the MAX15004/MAX15005 reduces the capacitance requirement.
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4.5V to 40V Input Automotive Flyback/Boost/SEPIC Power-Supply Controllers MAX15004/MAX15005
An additional small LC filter may be necessary to suppress the remaining low-energy high-frequency spikes. The LC filter also helps attenuate the switching frequency ripple. Care must be taken to avoid any compensation problems due to the insertion of the additional LC filter. Design the LC filter with a corner frequency at more than a decade higher than the estimated closed-loop, unity-gain bandwidth to minimize its effect on the phase margin. Use 1F to 10F low-ESR ceramic capacitors and calculate the inductance using following equation: L 1 4 x 10 x fc x C
3 2
above. Lower ripple current means lower peak and RMS currents and lower losses. The higher inductance value needed for a lower ripple current means a larger-sized inductor, which is a more expensive solution. The inductors L1 and L2 can be independent, however, winding them on the same core reduces the ripple currents. Calculate the maximum duty cycle using the following equation and choose the RT and CT values accordingly for a given switching frequency (see the Oscillator Frequency/External Synchronization section). VOUT + VD DMAX = VIN-MIN + VOUT + VD - (VDS + VCS ) where VD is the forward voltage of the Schottky diode, V CS (0.305V) is the current-sense threshold of the MAX15004/MAX15005, and VDS is the voltage drop across the switching MOSFET during the on-time.
where fC = estimated converter closed-loop unity-gain frequency.
SEPIC Converter
The MAX15004A/B/MAX15005A/B can be configured for SEPIC conversion when the output voltage must be lower and higher than the input voltage when the input voltage varies through the operating range. The dutycycle equation: VO D = VIN 1 - D indicates that the output voltage is lower than the input for a duty cycle lower than 0.5 while VOUT is higher than the input at a duty cycle higher than 0.5. The inherent advantage of the SEPIC topology over the boost converter is a complete isolation of the output from the source during a fault at the output. For the MAX15004/MAX15005, the SEPIC converter output can be fed back to VCC (Figure 6), so that the controller can function even during cold-crank input voltage ( 2.5V). Use a Schottky diode (DVIN) in the VIN path to avoid backfeeding the input source. A current-limiting resistor (RVCC) is also needed from the output to VCC depending upon the converter output voltage. The total VCC current sink must be limited to 25mA. See the Selecting VCC Resistor (RVCC) section to calculate the optimum value of the VCC resistor. The SEPIC converter design includes sizing of inductors, a MOSFET, series capacitance, and the rectifier diode. The inductance is determined by the allowable ripple current through all the components mentioned
Inductor Selection in SEPIC Converter Use the following equations to calculate the inductance values. Assume both L1 and L2 are equal and that the inductor ripple current (IL) is equal to 20% of the input current at nominal input voltage to calculate the inductance value.
V x DMAX L = L1 = L2 = IN-MIN 2 x fOUT x IL 0.2 x IOUT -MAX x DMAX IL = (1 - DMAX ) x where fOUT is the converter switching frequency and is the targeted system efficiency. Use the coupled inductors MSD-series from Coilcraft or PF0553-series from Pulse Engineering, Inc. Make sure the inductor saturating current rating (ISAT) is 30% higher than the peak inductor current calculated using the following equation. Use the current-sense resistor calculated based on the ILPK value from the equation below (see the Current Limit section). I x DMAX ILPK = OUT -MAX + IOUT -MAX + IL (1 - DMAX ) x
20
______________________________________________________________________________________
4.5V to 40V Input Automotive Flyback/Boost/SEPIC Power-Supply Controllers
MOSFET, Diode, and Series Capacitor Selection in a SEPIC Converter For the SEPIC configuration, choose an n-channel MOSFET with a VDS rating at least 20% higher than the sum of the output and input voltages. When operating at a high switching frequency, the gate charge and switching losses become significant. Use low gatecharge MOSFETs. The RMS current of the MOSFET is:
The series capacitor should be chosen for minimum ripple voltage (VCP) across the capacitor. We recommend using a maximum ripple VCP to be 5% of the minimum input voltage (VIN-MIN) when operating at the minimum input voltage. The multilayer ceramic capacitor X5R and X7R series are recommended due to their high ripple current capability and low ESR. Use the following equation to calculate the series capacitor CP value. I x DMAX CP = OUT -MAX VCP x fOUT where VCP is 0.05 x VIN-MIN. For a further discussion of SEPIC converters, go to http://pdfserv.maxim-ic.com/en/an/AN1051.pdf.
MAX15004/MAX15005
IMOS -RMS (A) =
[
D (ILPK ) + (ILDC ) + (ILPK x ILDC ) x MAX 3
2 2
]
where ILDC = (ILPK - IL). Use Schottky diodes for higher conversion efficiency. The reverse voltage rating of the Schottky diode must be higher than the sum of the maximum input voltage (VIN-MAX) and the output voltage. Since the average current flowing through the diode is equal to the output current, choose the diode with forward current rating of IOUT-MAX. The current sense (RS) can be calculated using the current-limit threshold (0.305V) of MAX15004/MAX15005 and ILPK. Use a diode with a forward current rating more than the maximum output current limit if the SEPIC converter needs to be output short-circuit protected. 0.305 RCS = ILPK Select R CS 20% below the value calculated above. Calculate the output current limit using the following equation: D IOUT -LIM = x (ILPK - IL ) (1 - D) where D is the duty cycle at the highest input voltage (VIN-MAX).
Power Dissipation
The MAX15004/MAX15005 maximum power dissipation depends on the thermal resistance from the die to the ambient environment and the ambient temperature. The thermal resistance depends on the device package, PCB copper area, other thermal mass, and airflow. Calculate the temperature rise of the die using following equation: TJ = TC + (PT x JC) or TJ = TA + (PT x JA) where JC is the junction-to-case thermal impedance (3C/W) of the 16-pin TSSOP-EP package and PT is power dissipated in the device. Solder the exposed pad of the package to a large copper area to spread heat through the board surface, minimizing the case-toambient thermal impedance. Measure the temperature of the copper area near the device (TC) at worst-case condition of power dissipation and use 3C/W as JC thermal impedance. The case-to-ambient thermal impedance (JA) is dependent on how well the heat is transferred from the PCB to the ambient. Use a large copper area to keep the PCB temperature low. The JA is 38C/W for TSSOP-16-EP and 90C/W for TSSOP-16 package with the condition specified by the JEDEC51 standard for a multilayer board.
______________________________________________________________________________________
21
4.5V to 40V Input Automotive Flyback/Boost/SEPIC Power-Supply Controllers MAX15004/MAX15005
VIN 2.5V TO 16V L1 L11 = L22 = 7.5H D2 STP745G D1 LL4148 R5 10 1 C1 100nF IN VCC 16 CVCC 1F VOUT D3 BAT54C C4 22F C1 6.8F C2 6.8F C3 6.8F C5 22F C6 22F VOUT (8V/2A)
C7 6.8F
MAX15005A/B
ON OFF 2 ON/OFF OUT 3 OVI 14 15 RG 1 STD20NF06L
PGND 4 CSLOPE 47pF REG5 5 6 SLOPE
REG5 13 C10 1F
REG5
RT 15k CT 150pF
N.C. RTCT CS 12 CCS 100pF
RCS 100 RS 0.025
7
SGND 11 R3 1.8k C3 47nF VOUT C4 680pF R2 15k
SYNC 8 RSYNC 10k SYNC
COMP
FB
10
SS
9 CSS 150nF
R1 2.7k
EP
Figure 6. SEPIC Application Circuit
22
______________________________________________________________________________________
4.5V to 40V Input Automotive Flyback/Boost/SEPIC Power-Supply Controllers
Layout Recommendations
Typically, there are two sources of noise emission in a switching power supply: high di/dt loops and high dv/dt surfaces. For example, traces that carry the drain current often form high di/dt loops. Similarly, the heatsink of the MOSFET connected to the device drain presents a dv/dt source; therefore, minimize the surface area of the heatsink as much as possible. Keep all PCB traces carrying switching currents as short as possible to minimize current loops. Use a ground plane for best results. Careful PCB layout is critical to achieve low switching losses and clean, stable operation. Refer to the MAX15005 EV kit data sheet for a specific layout example. Use a multilayer board whenever possible for better noise immunity. Follow these guidelines for good PCB layout: 1) Use a large copper plane under the package and solder it to the exposed pad. To effectively use this copper area as a heat exchanger between the PCB and ambient, expose this copper area on the top and bottom side of the PCB. 2) Do not connect the connection from SGND (pin 7) to the EP copper plane underneath the IC. Use midlayer-1 as an SGND plane when using a multilayer board. 3) Isolate the power components and high-current path from the sensitive analog circuitry. 4) Keep the high-current paths short, especially at the ground terminals. This practice is essential for stable, jitter-free operation. 5) Connect SGND and PGND together close to the device at the return terminal of VCC bypass capacitor. Do not connect them together anywhere else. 6) Keep the power traces and load connections short. This practice is essential for high efficiency. Use thick copper PCBs (2oz vs. 1oz) to enhance fullload efficiency. 7) Ensure that the feedback connection to FB is short and direct. 8) Route high-speed switching nodes away from the sensitive analog areas. Use an internal PCB layer for SGND as an EMI shield to keep radiated noise away from the device, feedback dividers, and analog bypass capacitors. 9) Connect SYNC pin to SGND when not used.
MAX15004/MAX15005
______________________________________________________________________________________
23
4.5V to 40V Input Automotive Flyback/Boost/SEPIC Power-Supply Controllers MAX15004/MAX15005
Typical Operating Circuits
VIN (5.5V TO 16V) R7 510 C12 220pF VANODE (110V/55mA) C13 10F 200V R8 100k D1 D2 VGRID (60V/12mA) C15 22F 60V C3 1F 16V R9 NU C14 NU R10 36k FILAMENT+ (3V/650mA) D4 ON/OFF R3 50 N FILAMENTC17 2.2F 10V D5 R15 100 C16 330F 6.3V
C1 330F 50V
R16 10 C18 4700pF 100V 1 C2 0.1F 50V 16
C11 2200pF 100V
R2 560
D2
IN
VCC
R17 100k 1% R18 47.5k 1%
MAX15005A/B
2 VIN
R11 182k 1% R12 12.1k 1%
OUT 3 OVI PGND
15
14
4 C4 100pF R1 8.45k 1% C5 1200pF 7 5 6
REG5 SLOPE REG5 13 C10 1F N.C. RTCT CS 12 C9 560pF R5 1k R6 0.06 1%
REG5
SGND COMP 11 R2 402k 1% 10 9 C8 0.1F C6 4700pF VANODE C7 47pF R13 118k 1% R14 1.3k 1%
8 R19 10k 1 JU1 2
SYNC
FB SS EP
Figure 7. VFD Flyback Application Circuit
24 ______________________________________________________________________________________
4.5V to 40V Input Automotive Flyback/Boost/SEPIC Power-Supply Controllers
Typical Operating Circuits (continued)
VIN (4.5V TO 16V)
MAX15004/MAX15005
C1 10F 25V
L1 10H/IHLP5050 VISHAY
1 C11 0.1F R11 301k 2 R10 100k R8 153k 3 R9 10k VOUT
IN
VCC
16 C10 1F/16V CERAMIC D1 B340LB VOUT (18V/2A) C6 56F/25V SVP-SANYO Q Si736DP
MAX15005A/B
ON/OFF R1 5
OUT OVI PGND
15
14
4 C2 100pF 5 6
REG5 SLOPE REG5 13 C10 1F N.C. RTCT CS 12 C4 100pF R3 1k R4 0.025
REG5
R2 13k C3 180pF
7
SGND COMP 11 R5 100k C9 0.1F VOUT C8 330pF R6 136k
SYNC 8 1 JU1 2 SYNC
FB SS EP
10 9
C7 0.1F
R7 10k
Figure 8. Boost Application Circuit
______________________________________________________________________________________ 25
4.5V to 40V Input Automotive Flyback/Boost/SEPIC Power-Supply Controllers MAX15004/MAX15005
Pin Configurations
TOP VIEW
IN 1 ON/OFF 2 OVI 3 SLOPE 4 N.C. 5 RTCT 6 SGND 7 SYNC 8
+
16 VCC 15 OUT 14 PGND IN 1 ON/OFF 2 OVI 3 SLOPE 4 N.C. 5 RTCT 6 SGND 7 SYNC 8
+
16 VCC 15 OUT 14 PGND
MAX15004A MAX15005A
13 REG5 12 CS 11 COMP 10 FB
MAX15004B MAX15005B
13 REG5 12 CS 11 COMP 10 FB 9 SS
EP*
9 SS
TSSOP-EP
*EP = EXPOSED PAD.
TSSOP
Chip Information
PROCESS: BiCMOS
Package Information
For the latest package outline information and land patterns, go to www.maxim-ic.com/packages. Note that a "+", "#", or "-" in the package code indicates RoHS status only. Package drawings may show a different suffix character, but the drawing pertains to the package regardless of RoHS status. PACKAGE TYPE 16 TSSOP 16 TSSOP-EP PACKAGE CODE U16+2 U16E+3 OUTLINE NO. 21-0066 21-0108 LAND PATTERN NO. 90-0117 90-0120
26
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4.5V to 40V Input Automotive Flyback/Boost/SEPIC Power-Supply Controllers
Revision History
REVISION NUMBER 0 1 2 REVISION DATE 1/07 11/07 12/10 Initial release Update Features, revise equations on pages 13, 20, and 21, revise Figure 8 with correct MOSFET, and update package outline Add MAX15005BAUE/V+ automotive part, update Features, update Package Information, style edits DESCRIPTION PAGES CHANGED -- 1, 13, 20, 21, 25, 28 1-5, 9, 13, 21, 25-29
MAX15004/MAX15005
Maxim cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim product. No circuit patent licenses are implied. Maxim reserves the right to change the circuitry and specifications without notice at any time.
Maxim Integrated Products, 120 San Gabriel Drive, Sunnyvale, CA 94086 408-737-7600 ____________________ 27
(c) 2010 Maxim Integrated Products Maxim is a registered trademark of Maxim Integrated Products, Inc.

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